1. Field of the Invention
The present invention relates to an apparatus and a method for reducing a peak-to-average power ratio (hereinafter, referred to as “PAPR”) in an Orthogonal Frequency Division Multiplexing (hereinafter, referred to as “OFDM”) communication system, and more particularly to an apparatus and a method for reducing PAPR by means of a complex gradient algorithm.
2. Description of the Related Art
Recently, a fourth generation (4G) mobile telecommunication system is being developed from the third generation (3G) mobile telecommunication system. The fourth generation (4G) mobile telecommunication system aims not only to provide the same mobile telecommunication services as those by the mobile telecommunication systems of previous generations but also to effectively interconnect a wire communication network and a wireless communication network and provide a service combining wire and wireless communication networks. Further, the 4G mobile telecommunication system arranges standards for technologies for providing a higher speed data transmission service than the 3G mobile telecommunication system.
Here, OFDM scheme is being actively studied for its application to the 4G mobile telecommunication system, and it employs a data transmission scheme using multi-carriers. The OFDM scheme is one multi-carrier modulation scheme in which symbols inputted in series are serial-to-parallel converted so as to be modulated into a plurality of sub-carriers having orthogonality to each other, i.e., a plurality of sub-carrier channels, which are then transmitted.
OFDM is similar to conventional frequency division multiplexing (FDM) but has a different characteristic in that a plurality of sub-carriers are transmitted while maintaining orthogonality between them in the OFDM scheme so that the OFDM scheme can achieve an optimum transmission efficiency. In other words, the OFDM scheme has good efficiency in using frequencies and is highly resistant to multi-path fading, so that the OFDM scheme can achieve optimum transmission efficiency in high speed data transmission.
Hereinafter, constructions of transmission/reception terminals of a communication system employing conventional OFDM will be described with reference to FIG. 1.
FIG. 1 is a block diagram showing constructions of transmission/reception terminals of a conventional OFDM mobile communication system.
Referring to FIG. 1, a mobile communication system using OFDM scheme includes a transmission terminal 100 and a reception terminal 150. The transmission terminal 100 includes a data transmitter 102, a coder 104, a symbol mapper 106, a serial to parallel (hereinafter, referred to as “S/P”) converter 108, a pilot symbol inserter 110, an inverse fast Fourier transform (hereinafter, referred to as “IFFT”) unit 112, a parallel to serial (hereinafter, referred to as “P/S”) converter 114, a guard interval inserter 116, a digital-to-analog converter (hereinafter, referred to as “D/A converter”) 118, and a radio frequency (hereinafter, referred to as “RF”) processor 120.
In the transmission terminal 100, the data transmitter 102 generates and outputs user data bits and control data bits to be transmitted to the coder 104. The coder 104 receives and codes the signals outputted from the data transmitter 102 according to a predetermined coding scheme, and then outputs the coded signals to the symbol mapper 106. The coder 104 may perform coding by means of a convolutional coding scheme or a turbo coding scheme having a predetermined coding rate. The symbol mapper 106 modulates the coded bits outputted from the coder 104 according to a corresponding modulation scheme, thereby generating modulation symbols, and outputs the modulation symbols to the S/P converter 108. Here, the modulation scheme that the symbol mapper 106 may follow includes, e.g., a BPSK (binary phase shift keying) scheme, a QPSK (quadrature phase shift keying) scheme, a 16QAM (quadrature amplitude modulation) scheme, 64QAM scheme, or others.
The S/P converter 108 receives and converts the serial modulation symbols outputted from the symbol mapper 106 into parallel modulation symbols, and outputs the converted parallel modulation symbols to the pilot symbol inserter 110. The pilot symbol inserter 110 inserts pilot symbols into the converted parallel modulation symbols outputted from the S/P converter 108 and then outputs them to the IFFT unit 112.
The IFFT unit 112 receives the signals outputted from the pilot symbol inserter 110, performs N-point IFFT for the signals, and then outputs them to the P/S converter 114.
The P/S converter 114 receives the signals outputted from the IFFT unit 112, converts the signals into serial signals, and outputs the converted serial signals to the guard interval inserter 116. The guard interval inserter 116 receives the signals outputted from the P/S converter 114, inserts guard intervals into the received signals, and then outputs them to the D/A converter 118. Here, the inserted guard interval prevents interference between OFDM symbols transmitted in the OFDM communication system; that is to say, the inserted guard interval prevents interference between a previous OFDM symbol transmitted during a previous OFDM symbol period and a current OFDM symbol to be transmitted during a current OFDM symbol period.
In inserting the guard interval, a method of inserting null data has been proposed. However, insertion of null data as a guard interval may cause a receiver to erroneously estimate a start point of an OFDM symbol, thereby allowing interference between sub-carriers, which increases the probability of erroneous determination of the starting point for received OFDM symbols. Therefore, a cyclic prefix method or a cyclic postfix method is usually used in inserting the guard interval. In the cyclic prefix method, a predetermined number of last bits of an OFDM symbol are copied and inserted into an OFDM symbol. In the cyclic postfix method, a predetermined number of initial bits of an OFDM symbol are copied and inserted into an OFDM symbol.
The D/A converter 118 receives the signals outputted from the guard interval inserter 116, converts the signals into analog signals, and outputs the converted analog signals to the RF processor 120. The RF processor 120 includes a filter and a front end unit. The RF processor 120 receives the signals from the D/A converter 118, RF-processes the signals, and then transmits the signals over the air through a Tx antenna.
Hereinafter, the reception terminal 150 will be described.
The reception terminal 150 includes an RF processor 152, an analog-to-digital converter (hereinafter, referred to as “A/D converter”) 154, a guard interval remover 156, a S/P converter 158, a fast Fourier transform (hereinafter, referred to as “FFT”) unit 160, a pilot symbol extractor 162, a channel estimator 164, an equalizer 166, a P/S converter 168, a symbol demapper 170, a decoder 172, and a data receiver 174.
The signals transmitted from the transmission terminal 100 pass through multi-path channels and are received by an Rx antenna of the reception terminal 150 in a state in which noise is included in the signals. The signals received through the Rx antenna are inputted to the RF processor 152, and the RF processor 152 down-converts the received signals into signals of an intermediate frequency (IF) band, and then outputs the IF signals to the A/D converter 154. The A/D converter 154 converts the analog signals outputted from the RF processor 152 into digital signals and then outputs the digital signals to the guard interval remover 156.
The guard interval remover 156 receives the digital signals converted by and outputted from the A/D converter 154, eliminates guard intervals from the digital signals, and then outputs them to the S/P converter 158. The S/P converter 158 receives the serial signals outputted from the guard interval remover 156, converts the serial signals into parallel signals, and then outputs the parallel signals to the FFT unit 160. The FFT unit 160 performs N-point FFT on the signals outputted from the P/S converter 158, and then outputs them to both the equalizer 166 and the pilot symbol extractor 162. The equalizer 166 receives the signals from the FFT unit 160, channel-equalizes the signals, and then outputs the channel-equalized signals to the P/S converter 168. The P/S converter 168 receives the parallel signals outputted from the equalizer 166, converts the parallel signals into serial signals, and then outputs the converted serial signals to the symbol demapper 170.
As indicated, the signals outputted from the FFT unit 160 are also inputted to the pilot symbol extractor 162. The pilot symbol extractor 162 detects pilot symbols from the signals outputted from the FFT unit 160 and outputs the detected pilot symbols to the channel estimator 164. The channel estimator 164 performs channel estimation using the pilot symbols and outputs the result of the channel estimation to the equalizer 166. Here, the reception terminal 150 generates channel quality information (hereinafter, referred to as “CQI”) corresponding to the result of the channel estimation and transmits the CQI to the transmission terminal 100 through a CQI transmitter (not shown).
The symbol demapper 170 receives the signals outputted from the P/S converter 168, demodulates the signals according to a demodulation scheme corresponding to the modulation scheme of the transmission terminal 100, and then outputs the demodulated signals to the decoder 172. The decoder 172 decodes the signals from the symbol demapper 170 according to a decoding scheme corresponding to the coding scheme of the transmission terminal 100 and outputs the decoded signals to the data receiver 174.
However, the OFDM system not only has the advantages described above but is also problematic in that the multi-carrier modulation may cause a high PAPR in the OFDM system. That is, in the OFDM system, data is transmitted by means of multiple carriers, so that a resultant OFDM signal has an amplitude, which is equivalent to the sum of the amplitudes of all the carriers, and thus may largely change. In particular, when the multiple carriers have the same phase, the amplitude of the resultant OFDM signal may have a very large value and may fluctuate greatly. Therefore, the OFDM signal may go beyond an operation range of a high power linear amplifier (not shown) and thus may be distorted after passing the high power linear amplifier. In order to prevent this distortion, the high power linear amplifier employs a back-off method for allowing the signal to be maintained within the linear range by lowering an input power.
That is to say, in the back-off method, the operation point of the high power linear amplifier is lowered in order to reduce distortion of the signal. However, the larger the back off value is, the more inefficient the utilization of the amplifier is. Therefore, a signal having a high PAPR may deteriorate the efficiency of the linear amplifier.
Typical methods for reducing a PAPR in an OFDM communication system include clipping, block coding, phase adjustment, and tone reservation (hereinafter, referred to as “TR”).
In the clipping method, in order to allow a signal to have an amplitude within a linear operation range of an amplifier, when the amplitude of the signal exceeds a predetermined reference clipping value set in advance, a portion of the amplitude of the signal exceeding the reference clipping value is clipped out. However, in the clipping method, non-linear operation may cause in-band distortion, thereby increasing inter-symbol interference and bit error rate. Further, in the clipping method, out-band noise may cause channel interference, thereby deteriorating spectrum efficiency.
In the block coding method, extra carriers are coded and then transmitted, in order to lower the PAPR of all the carrier signals. In this method, the coding of the extra carriers achieves the correction of errors and reduction of the PAPR without distortion of the signal. However, when sub-carriers have large amplitudes, this method yields a very bad spectrum efficiency and requires a large look-up table or a large generation matrix, increasing and greatly complicating calculations.
The phase adjustment method includes a partial transmit sequence (hereinafter, referred to as “PTS”) method and a selective mapping (hereinafter, referred to as “SLM”) method.
In the PTS method, input data is divided into M sub-blocks, each of the M sub-blocks is subjected to L-point IFFT and is then multiplied by a phase factor for minimizing the PAPR, and then the M sub-blocks are summed and transmitted.
In the SLM method, the same M data blocks are multiplied by different phase sequences having statistically independent N lengths, and one of the multiplied blocks, which has the lowest PAPR, is selected and transmitted. The SLM method requires M IFFT procedures but can considerably lower the PAPR and can be applied to all the carriers regardless of the number of the carriers.
However, both the PTS method and the SLM method are problematic in that additional information on rotation factors must be transmitted to a receiver in order to restore data. Such a transmission of the additional information complicates the communication method and causes all OFDM symbol information within a corresponding period containing an erroneous symbol to be treated as erroneous.
Meanwhile, in the TR method, some tones from among the entire sub-carriers, which carry no data, are reserved for PAPR reduction. Here, the receiver disregards the tones carrying no information signal and restores information signals from the other tones. Therefore, the receiver may have a simpler construction.
The gradient algorithm is a good solution for the TR method. In the gradient algorithm, which is an application of the clipping method to the TR method, signals having an impulse characteristic are generated using the tones carrying no information signal, and IFFT output signals are clipped using the signals having the impulse characteristic. When the generated signals having an impulse characteristic are added to the IFFT output signals, data distortion occurs only in some tones carrying no information but does not occur in the other tones of frequency domain.
Hereinafter, the TR method using the gradient algorithm will be described with reference to FIG. 2.
FIG. 2 illustrates a construction of a transmitter using the conventional TR method.
Referring to FIG. 2, a total of N sub-carriers outputted through the S/P converter 108 of FIG. 1 include L tone signals 201 and (N-L) information signals 203. Here, the information signals refer to user data bits and control data bits. Further, the L reserved tone signals carrying no information produce a waveform having an impulse characteristic and are used to clip the output signals of the IFFT unit 112.
The (N-L) number of the information signals 203 and the L number of the reserved tone signals 201 are inputted to a tone allocation unit 205. The tone allocation unit 205 allocates the L reserved tone signals 201 to positions of sub-carriers reserved in advance between the transmitter and the receiver. In other words, the tone allocation unit 205 allocates the L reserved tone signals 201 to reserved L positions, allocates (N-L) tones or (N-L) information signals 203 to the remaining (N-L) positions, and then outputs them to an N-point IFFT unit 207.
The N-point IFFT unit 207 receives all allocated tone signals, performs an IFFT operation on them, and then outputs them to a parallel-serial converter 209. The parallel-serial converter 209 receives the parallel signals having been subjected to the IFFT operation, converts the parallel signals into serial signals, and then outputs the converted serial signals to a gradient algorithm unit 211. Here, when the converted serial signals are assumed to be x, x represents signals of the time domain. The gradient algorithm unit 211 generates a signal c of the time domain, adds the signals c and x, and then outputs a transmission signal, which is the sum of the signals c and x.
Herein, the signal c used for reduction of the PAPR can be expressed as Equation 1.
                              C          k                =                  {                                                                                          C                    k                                    ,                                                                              k                  ∈                                      {                                                                  i                        1                                            ,                                              i                        2                                            ,                      ⋯                      ⁢                                                                                          ,                                              i                        L                                                              }                                                                                                                        0                  ,                                                                              k                  ∉                                      {                                                                  i                        1                                            ,                                              i                        2                                            ,                      ⋯                      ⁢                                                                                          ,                                              i                        L                                                              }                                                                                                          (        1        )            
In Equation 1, L sub-carriers are reserved in advance and used for the signal C, and the locations {i1 . . . , iL} of the L sub-carriers are fixed by the tone allocation unit 205 at the time of initial transmission. Further, in Equation 1, i represents indices of reserved tone signals in the tone allocation unit 205 and k represents indices of frequency domain. Herein, input signals X are allocated to the sub-carriers other than signals c in the manner as expressed by Equation 2.
                              X          k                =                  {                                                                                          X                    k                                    ,                                                                              k                  ∉                                      {                                                                  i                        1                                            ,                                              i                        2                                            ,                      ⋯                      ⁢                                                                                          ,                                              i                        L                                                              }                                                                                                                        0                  ,                                                                              k                  ∈                                      {                                                                  i                        1                                            ,                                              i                        2                                            ,                      ⋯                      ⁢                                                                                          ,                                                                                          ⁢                                              i                        L                                                              }                                                                                                          (        2        )            
Hereinafter, the conventional gradient algorithm as described above will be described in detail with reference to FIG. 3, which is a block diagram of an apparatus for reducing PAPR using the conventional gradient algorithm.
Referring to FIG. 3, an apparatus for reducing PAPR using the conventional gradient algorithm includes a p waveform generator 301, a peak detector 303, a circular shift unit 305, a scaling unit 307, an adder 309, a PAPR calculation unit 311, and a control unit 313.
First, the p waveform generator 301 generates a p waveform from the L tones 201, positions of which have been reserved in the tone allocation unit 205, from among the total N signals. The p waveform is a signal similar to an impulse signal, which has been obtained through several hundred thousand times to several million times of repetitive random selections of at least one tone having no information from among the entire signals.
Meanwhile, the serial signals x of the time domain, which have been converted from analog signals after being subjected to IFFT, are inputted to the gradient algorithm unit 211. The peak detector 303 detects the maximum peak value and the peak position of the signals x inputted to the gradient algorithm unit 211. The circular shift unit 305 circularly shifts the location of the p waveform to the detected maximum peak position. The scaling unit 307 scales the maximum peak value of the signals x with the circularly shifted p waveform so that the maximum peak value of the signals x outputted after being subjected to IFFT can be maintained below a PAPR value set in advance by the system. Here, if a scaled value for lowering the maximum peak value below the predetermined PAPR is c, it can be said that c is an optimum value calculated by the gradient algorithm unit 211 in order to eliminate the peak value of the output signal x of the N-point IFFT unit 207.
The adder 309 adds the signals x and c and outputs the sum to the PAPR calculation unit 311. The PAPR calculation unit 311 calculates a PAPR for the inputted signal x+c and transmits the calculated value to the control unit 313.
The control unit 313 receives the calculated PAPR value, and feedbacks the PAPR value and repeatedly performs the gradient algorithm when the calculated PAPR value is larger than the PAPR value set in the system. Such feedback of the PAPR value and re-execution of the gradient algorithm are repeated until the calculated PAPR value becomes smaller than the PAPR value set in the system. However, in order to prevent infinite repetition, the system has a preset maximum limit for the number of times of repetition and transmits the signal when the gradient algorithm has been repeatedly executed by the preset times even though the calculated PAPR is larger than the PAPR set in the system.
When it is assumed that an N-point IFFT output value is a complex number a+bi (where i=√{square root over (−1)}), the conventional gradient algorithm takes only the real number term into account (b=0). However, it is preferred that the OFDM system uses all sub-carriers for high speed data transmission. When all the sub-carriers are used, the IFFT output has a value of a complex number, and it is thus impossible to employ the conventional gradient algorithm controlling the amplitude of the signal only for the real number term.
Further, in the gradient algorithm, in order to enable the IFFT output signal to have a real number, the input data in the frequency domain must be symmetrical and conjugate. When the outputs after the IFFT in the frequency domain are symmetrical and conjugate, they have the same amplitude and a phase difference of 180°, so that the imaginary number term b becomes zero. Therefore, the conjugate portions cannot carry data, and thus throughput of the system is lowered to one-half capacity. Therefore, the conventional gradient algorithm reflecting only the real number term has a low throughput and is thus inefficient for high speed data transmission.